Hybrid filter for high slew rate output current application

ABSTRACT

An active linear regulator circuit in parallel with a filter capacitor of a switching voltage regulator injects current to a load only when the switching regulator and capacitor cannot supply adequate current to follow high frequency load transients in a manner which is compatible with adaptive voltage positioning (AVP) requirements. control of current injection and determination of the insufficiency of current from the switching regulator and capacitors is achieved by impedance matching of the linear regulator to the switching regulator. The linear regulator thus operates at relatively low current and duty cycle to limit power dissipation therein. By matching impedances and increasing the bandwidth of the switching regulator, filter capacitor requirements can be reduced to the point of being met entirely by packaging and/or on-die capacitors which may be placed close to or at the point of load to reduce parasitic inductance, as can the linear regulator.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention generally relates to power supplies for electronicdevices and, more particularly, high current, low voltage loads ofcomplex electronic devices such as computers and telecommunicationsdevices which operate at high frequencies.

2. Description of the Prior Art

Recent developments in many types of electronic devices have causedpower supply requirements to become much more stringent to the pointthat current power supply designs are becoming inadequate orimpractical. For example, switching power supplies are well known forDC-DC conversion and operate by switching current from a relatively highvoltage source which is then filtered, often using a series inductance,to maintain an approximate desired voltage on a bulk capacitor, oftenreferred to as a filter capacitor. Switching frequency and/or duty cycleto control output voltage is determined by feedback of variation involtage in a known manner. However, such feedback involves anunavoidable delay in the response to transient changes in load andlimits accuracy of voltage regulation.

Thus, power supplies of this and other known types rely on the largecapacitance value of the bulk or filter capacitor to supply potentiallylarge transient currents. Such filtering with capacitors may be requiredin several stages (e.g. following the bulk capacitor) of a powerdistribution network where the network may branch or present arelatively long connection (such capacitors being referred to asdecoupling capacitors) as well as at the point of load (where suchcapacitors are often physically located on the back of a CPU andreferred to as packaging capacitors) such as at an integrated circuitpackage or even at various locations on an integrated circuit chip tominimize effects of voltage ripple and variation due to load transientsand propagation thereof through the power distribution network from oneportion of a circuit or device to another. The slew rate of currentchange is also compromised by the total capacitance presented by thesefilter and decoupling capacitors. Such effects are particularly evidentin regard to current and foreseeable complex digital circuits such asmicroprocessors or telecommunications devices which operate at extremeclock speeds and where the load can change radically from one clockpulse to the next and parasitic inductance and capacitance of powerconnections may cause large changes in impedance of the power supplyconnections.

Moreover, the trend toward increased complexity and reduced size ofintegrated circuit chips has resulted in designs which operate atrelatively lower voltages (e.g. less than 1V) and higher currents (e.g.150 A or higher—also aggravated by increased numbers of active devicessuch as transistors formed on each chip) supplied through connectionspresenting potentially significant parasitic impedances, particularly asthe trend toward higher clock frequencies continues. Further, as amatter of thermal management, some integrated circuits, notablymicroprocessors, specify a variation of supply voltage with load whichis referred to as adjustable voltage positioning (AVP). Therefore,voltage regulation tolerance has become smaller and more stringent aswell as more complex while the potential magnitude of current transientsand voltage drops in power connections increases. It is projected thatthe capacitance requirements for bulk capacitors will increase by afactor of 2.5 for bulk capacitors, a factor of 13 for decouplingcapacitors outside a device connection (e.g. at an IC socket or board)and a factor of 6 for so-called packaging capacitors within or closelyassociated with an IC package or on-chip to derive sufficient voltageregulation performance using current voltage regulator and filter designto meet current and foreseeable voltage regulation requirements. Suchlarge increases in required capacitance can be better appreciated fromthe fact that increased filter and decoupling capacitance compromisesslew rate and bandwidth of the voltage regulator due to increasingnumbers of output capacitors for energy storage, thus requiring evengreater capacitance to provide a given voltage regulation tolerancewhich is becoming more stringent, as alluded to above. Further, whilenumerous capacitor technologies are known for different ranges ofcapacitance value, required increases in capacitance of individualcapacitors is invariably accompanied by increases in physical size ofcapacitors and larger capacitance values must often be implemented intechnologies which require greater volume per unit of capacitance.

Other approaches to answering a need for increased current slew rate andimproved transient response are summarized in an article entitled“Hybrid Power filter with Output Impedance Control” by Y. Ren et al.,36^(th) Conference on Power Electronics Specialists, pp. 1434-1440, Jun.12, 2005, of which the inventors are co-authors and which is herebyfully incorporated by reference. However, each of these approaches hasengendered corresponding problems and/or limitations rendering theminadequate to the requirements for providing power to currentmicroprocessor and communications ICs. For example, a hybrid powersupply noted therein provides a linear source in parallel with aswitching regulator with the voltage loop closed through the linearsource and the current from the linear source used as an error signalfor the switching regulator. However, such an arrangement is inherentlyincompatible with adaptive voltage positioning (AVP) specified bymanufacturers of microprocessors and other ICs of high functionality.

SUMMARY OF THE INVENTION

It is therefore an object of the present invention to provide a voltageregulator and power distribution system of increased bandwidth; allowingreduction of output capacitors in capacitance value and size.

It is another object of the present invention to provide a hybrid powersupply filter arrangement which improves slew rate of change in currentoutput and presents a reduced impedance to loads.

In order to accomplish these and other objects of the invention, a powersupply and distribution network is provided comprising a voltageregulator including an output capacitor functioning as a current source,and a linear regulator functioning as a current source in parallel withthe output capacitor.

In accordance with another aspect of the invention, a method is providedfor reducing capacitance required for filtering output of a voltageregulator and/or increasing effective slew rate of said voltageregulator, comprising steps of setting bandwidth of the voltageregulator in accordance with a clock frequency of a load circuitreceiving power from the voltage regulator, injecting current into apower supply connection when required power transients exceed a currentslew rate of the voltage regulator.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing and other objects, aspects and advantages will be betterunderstood from the following detailed description of a preferredembodiment of the invention with reference to the drawings, in which:

FIG. 1 is a graph of historical and projected voltage and currentrequirements for microprocessors,

FIG. 2 is a graph of historical and projected values of slew rate andoutput impedance of power supplies and distribution networks of knowndesign,

FIG. 3 is a schematic representation of a power supply and distributionnetwork (with respect to a single point of load) of current design whichwill be useful in conveying an understanding of the principles of thepresent invention,

FIGS. 4 and 5 are graphical depictions of the relationship betweenvoltage regulator bandwidth and of the bulk capacitance and decouplingcapacitance, respectively,

FIG. 6 is a schematic depiction of a power supply and distributionnetwork including the hybrid filter in accordance with the invention,

FIG. 7 illustrates the conceptual waveform of the hybrid filter inaccordance with the invention,

FIG. 8 illustrates a worst case load transient,

FIG. 9 illustrates derivation of desired linear regulator waveforms inregard to the load transient of FIG. 8 in accordance with FIG. 7,

FIG. 10 illustrates conversion of the desired linear regulator waveformof FIG. 9 from the time domain to the frequency domain,

FIGS. 11 and 12 are frequency domain graphs of the output impedance ofthe switching regulator and the linear regulator, respectively, andapproximations thereof for practical implementation of the invention,

FIGS. 13 and 14 are schematic depictions of preferred forms of thecompensator and the power amplifier, respectively, for the linearregulator in accordance with the invention,

FIG. 13A is a graph of output impedance as a function of frequency ofthe switching regulator, capacitors and the linear regulator,

FIG. 15 is a schematic diagram of a preferred implementation of theinvention corresponding to the more generalized depiction of FIG. 6,

FIG. 16 shows a preferred implementation of the circuits of FIGS. 13 and14 to form a preferred embodiment of the linear regulator of FIG. 15,

FIG. 17 illustrates simulated performance of the invention for the worstcase load transient of FIG. 8, and

FIG. 18 is a side or cross-sectional view of a preferred form of powerdelivery structure for application of the invention.

DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT OF THE INVENTION

Referring now to the drawings, and more particularly to FIGS. 1 and 2,there is shown historical and projected future power supply current andvoltage requirements and slew rate and output impedance for powersupplies of current design for supplying power to microprocessors,respectively. As alluded to above, increases in integration density andreduction of minimum feature size regimes required to provideimprovements in performance and functionality of microprocessors inrecent years has led to a decrease in power supply voltage (e.g. toimprove breakdown margins and to reduce voltage swing to reduce currentsto capacitive elements such as gates of field effect transistors) fromabout 1.2 volts is 2001-2002 to about 1.0 volts at the present time witha further reduction to about 0.8 volts projected within the near future.Such a reduction not only reduces the voltage regulation tolerance (e.g.10% of rated voltage) but imposes even more stringent requirements forvoltage regulation accuracy. Over the same period, current requirementshave substantially doubled from 50 to 100 Amperes with a furtherincrease to 150 Amperes being projected as being required within a fewyears.

As alluded to above, increased filtering and decoupling capacitance aswell as the number of stages of filtering and decoupling and the numberof capacitors compromises slew rate of power supplies and it can be seenfrom FIG. 2 that the slew rate has increased from about 10Amperes/nanosecond to over 120 Amperes/nanosecond while the clock speedof microprocessors and other integrated circuits has been increasingsignificantly. Further, the clock speed is largely determinative of therate and magnitude of maximum load current change which the power supplyand power distribution network must accommodate. Thus, higherfrequencies present in the load current waveform increase the impedanceof the power supply and power distribution system as seen by the load,due largely to the parasitic inductances and capacitances thereof and itcan be seen from FIG. 2 that output impedance has, indeed, decreasedfrom about 2.5 milliohms in 2002 to about 1.0 milliohms at present witha further decrease to about 0.8 milliohms being projected within thenear future. These predictions carry a high degree of confidence sincethe degree of integration density and nominal clock frequencies arealready known for designs which exist currently and which will beimplemented at a commercial level at the projected times indicated.

FIG. 3 schematically illustrates an equivalent circuit of a projectedfuture switching regulator and power distribution network (with respectto a single point-of-load, for clarity) necessary to provide the voltageregulation performance to meet the requirements of foreseeablemicroprocessor and telecommunication ICs. Exemplary parasitic impedancesat particular points of the power supply and distribution network areillustrated. For reference, the circuit to the right of point/node A inFIG. 3 represents, for example, an integrated circuit such as amicroprocessor with an internal or even on-chip filter/decouplingcapacitor (referred to as a packaging capacitor) which may beimplemented in a distributed manner as a plurality of small (packaging)capacitors. (For example, the notation 1 μf*150 denotes 150 capacitorseach having a value of 1 μf. or the equivalent capacitance, differentlydistributed) The inductance illustrated between points A and B is theparasitic inductance of the connection from node B, where a decouplingcapacitor or a plurality thereof can be provided, to the chip. It shouldbe noted that a decoupling capacitor at point B has only recently beenconsidered necessary but is often provided at the present time,particularly as the impedance between points/nodes B and C has becomesignificant at higher IC clock frequencies. Further, such decouplingcapacitors are preferably provided at each power input pin of the ICwhich may number in the hundreds (or at groups thereof) to be as closeas possible to the load.

Point/node C, also commonly referred to as the sensing point,corresponds to the termination of the power distribution network inaccordance with prior designs and, topologically, would represent apoint “downstream” from a point at which a power distribution networkmight branch and, in any case, is often relatively remote from theswitching regulator (at and to the left of point/node D in FIG. 3) andthus the connection from point/node D to point/node C has relativelylarge and significant parasitic inductance and inherent resistance, asillustrated. Therefore, it has been the practice to provide a decouplingcapacitor or a plurality thereof at such a point at respective loads toimprove regulation and voltage stability and to prevent changes in loadin one portion of the power distribution circuit or board from affectingother devices and circuits being powered from the same switchingregulator.

Thus, the power distribution arrangement of FIG. 3 has at least threeloops, identified in FIG. 3 by curved arrows. Each of these loops willexhibit resonance at some frequency and thus be an additional potentialsource of variation in the voltage which can be output therefrom as wellas exhibiting a frequency dependent impedance which may limit thecurrent which may be delivered to a following stage or a load.Accordingly, to meet more stringent requirements in regard to voltageregulation tolerances and to do so at high frequencies, the above-notedincreases in capacitance (e.g. by a factor of 2.5 at point/node A, afactor of 13 at point/node B and a factor of 6 at point/node C) would berequired. Such increases in capacitance are not economically feasibleand would require more volume than is acceptable or, in some cases,physically possible while the potential for increase in parasitics ofmore numerous capacitors limit transient performance and efficiency.Moreover, as alluded to above, such increases in capacitance alsocompromise the bandwidth of the voltage regulator.

To provide a solution, the invention provides for increasing thebandwidth of the voltage regulator to provide for reduction orelimination of bulk/filter and decoupling capacitors by use of a hybridfilter design that includes active circuits, controlled by matching ofimpedances, capable of injecting current at the regulated voltage andwhich is conducive to the possibility of doing so in close physicalproximity to the point of load. The basic principles of operation andimplementation of the invention will now be explained with reference toFIGS. 4-12.

FIG. 4 is a graphic depiction of the relationship between the requiredvalue of the output capacitance, also referred to above as a bulk orfilter capacitance, at point/node D and control bandwidth, f_(c). Theincreasingly negative slope of the required filter capacitance valuewith increasing frequency indicates that smaller filter capacitorsprovide sufficient regulation as switching frequency is increased. Inother words, FIG. 4 graphically illustrates the trade-off betweenvoltage regulator switching frequency and the required capacitance valuefor a given voltage regulation tolerance which reflects the fact that ahigher switching frequency allows the switching regulator to moreclosely follow a load transient and directly supply a greater portion ofthe required transient current while the filter capacitor suppliescorrespondingly less.

Assuming a voltage regulator bandwidth target of 200 KHz and that thecontrol bandwidth, f_(c)=1/6*f_(s), the switching frequency, a voltageregulator switching frequency of 1.2 MHZ would be required. (The ratioof 1/6 is based on the control bandwidth necessary to avoid resonancebetween the voltage regulator module (VRM) inductor and capacitor inwhich the capacitor and inductor values are based on switchingfrequency. The ratio of 1/6 is valid for so-called buck step-downconverters and different ratios may be valid for other types ofconverters.) At this target switching frequency/bandwidth, thetechnology of the filter capacitor should be changed from Oscon (a typeof specialty polymer and aluminum electrolyte capacitor) to ceramicsince the latter is more cost effective and has a smaller footprint forthe required capacitance value. (For purposes of the present invention,the difference between Oscon and ceramic capacitors is a function of therespective per-piece capacitance, resistance and inductance. Both typesof capacitor have a slope of approximately −2 at low frequencies. Osconcapacitors have a much lower Q-factor and the slope becomes morenegative as compared with ceramic capacitors as frequency/bandwidth areincreased.) Also, the capacitor value sufficient to provide a givendegree of regulation varies as a function of the internal resistance ofthe capacitor and a reduced internal resistance allows a smallercapacitance to be employed, as depicted by the arrow at f_(c)=200 KHzbandwidth. That is, the target bandwidth of f_(c)=200 KHz allows asignificant reduction in required capacitance and space requiredtherefor and augmentation of that effect by substitution of capacitortechnology.

At a second voltage regulator bandwidth target of 350 KHz where therequired filter capacitance decreases markedly with controlfrequency/bandwidth increase, a voltage regulator switching frequency of2.0 MHZ is required. However, as indicated in FIG. 4, at such abandwidth, the bulk/filter/output capacitor(s) can be completelyeliminated since sufficient filtering can be achieved by capacitors(e.g. cavity, packaging, etc. capacitors) further “downstream”. That is,at this point, downstream capacitors are sufficient to handlehigher-frequency transients.

Similarly, FIG. 5 illustrates a generally similar relationship betweenrequired capacitance of decoupling capacitors (e.g. at point/node C),particularly in the vicinity of a third bandwidth target of 650 KHz anda voltage regulator switching frequency of 4.0 MHZ which would beconsidered to be extremely aggressive design, if reliably achievable atall (since parasitic inductance is difficult to control and may not beaccurately determinable). The solid portion of the top curve is based onthe current power delivery structure and shows that, at this frequencyand bandwidth, the number of decoupling capacitors can be reduced innumber from two hundred thirty to only fifty. However, beyond such afrequency/bandwidth, because the equivalent series inductance (e.g. ofthe power supply connection(s) and the decoupling capacitors), referredto as ESL (which causes unwanted high impedance at very high frequencieswhich the VRM, being upstream from the decoupling capacitors, cannotovercome), and interconnection parasitics dramatically hinders thecapacitance reduction as the bandwidth increases, it is not necessary topush the bandwidth higher in regard to current power deliverystructures. In other words, at higher switching frequencies/bandwidth,the parasitic inductances increases impedance such that no furtherincrease in supply of power directly from the VRM can be achieved andthe remaining amount of required power which must be supplied from thedecoupling capacitors cannot be further significantly reduced eventhough the parasitic inductance of the coupling capacitor(s) maycontribute to the ESL while ESL may be reduced by increase ofcapacitance.

If, however, the parasitic inductances can be reduced by relocation ofthe voltage regulator from the motherboard to an organic land grid array(OLGA) which is a type of low impedance substrate, as shown in FIG. 17,the number of decoupling capacitors can be reduced to 14*47 μF at the650 KHz bandwidth, as shown in FIG. 5. Further, as shown in FIG. 5, asuitable capacitance value can also be distributed to 360*1 μF packagingcapacitors (e.g. capacitors generally located beneath the microprocessorand/or attached to the bottom of the microprocessor package, asreflected in such nomenclature). That is, the required total capacitancevalue can be further reduced by placing the capacitors closer to theload thus effectively allowing omission of decoupling capacitors andperforming their filtering and current supply function with packagingcapacitors.

The linear regulator in accordance with the invention, as will bedescribed below, and preferably placed electrically between thepackaging capacitors and the microprocessor may replace or allowomission of many packaging capacitors but, in general, it is preferredto use as many packaging capacitors as space or other constraints allow,typically twenty-three to twenty-five, which may thus be used directlyas output capacitors. The reduction of capacitance which may be attainedis due to the much smaller ESL (since the ESL is reduced as the quantityof capacitance is increased). In order to further reduce the number ofpackaging capacitors to a number which may be conveniently accommodated,use of a linear regulator rather than a switching regulator (which islimited by f_(s)) has been proposed in order to answer the stringenttransient response requirement. A linear regulator provides theadvantage of being able to use a higher input voltage and, therefore, ahigher slew rate can be obtained. However, the major drawback of the useof linear regulators is their severe power dissipation; reduction ofwhich has proven intractable prior to the present invention.

The basic concept of the invention is to use a linear regulator only toaugment passive devices (e.g. capacitors) and to do so as part of a planto reduce undesired impedances by placing the VRM, capacitors and linearregulator closer to the microprocessor or other load. By controlling theoutput impedance at the linear regulator, it can be used only at veryhigh frequencies and lower losses may thus be achieved because thelinear regulator is used only at higher harmonics of any transients. Bythe same token, controlling the linear regulator through impedancematching rather than through any control signal allows instantaneousresponse without signal propagation delay and assures that the linearregulator will supply current only when the much more efficientcapacitors cannot fully follow and provide current corresponding to aload transient; thus minimizing current provided by the linear regulatorto maximize efficiency thereof.

FIG. 6 schematically illustrates the hybrid filter concept in accordancewith the invention. In comparison with the current approach to powersupply and power distribution network design of FIG. 3, it will beimmediately noted that no bulk/filter capacitor is included in theswitching regulator 60 and the packaging capacitor(s) 62 is useddirectly as the output capacitor, as alluded to above. A linearregulator 64 is placed in parallel with the output/packaging capacitor62 and both serve as current sources for the load 68 to accommodatecurrent transient requirements, as does the on-die capacitance 66 aswill now be explained with reference to FIG. 7.

The upper trace of FIG. 7 represents a current pulse i_(o) of idealizedform with the leading edge 70 of the pulse representing the requiredslew rate of the power supply and distribution network imposed by theload. Within this current pulse, line 72 represents the low slew rate ofthe switching regulator and the area below line 72 represents the energysupplied from the switching regulator 60 of FIG. 6. Line 74 representsthe time constant of packaging capacitor 62 with its series parasiticinductance and inherent resistance as illustrated in FIG. 6 and the areabetween lines 72 and 74 represents the energy transferred from thepackaging capacitor during the current pulse. Similarly, line 70represents the time constant of the on-die capacitor 66 (since theinductance and resistance of an on-die capacitor is extremely small)while area between line 70 and line 76 corresponds to the amount ofenergy which can be transferred therefrom during the pulse. Theremaining area between lines 74 and 76 thus represents the energy whichis provided from the linear regulator 64. The dashed lines in the lowerportion of FIG. 7 is a plot of the current required by the load otherthan that from the switching regulator 60 and on-die capacitor 66 whereline 76′ represents the required slew rate SR1. Solid line 74′represents the same area as that between lines 72 and 74 discussed aboveand corresponding to the energy transferred from the packagingcapacitor. It can be seen from a comparison of these areas that SR2<<SR1and is much less stringent because of the injection of current fromlinear regulator 64. By the same token, the amount of current and energyprovided by the linear regulator is relatively small and occurs duringonly a short portion of the current pulse (as will be even more evidentfrom FIG. 9, as will be discussed below). Therefore, even though linearregulators characteristically exhibit severe power dissipation, thetotal power dissipated by the linear regulator placed in a hybrid filterin accordance with the invention can be reduced to an acceptably lowlevel while the duty cycle of current injection is also relatively lowwhich facilitates thermal management. Thus the provision of a linearregulator as a current injector can achieve beneficial effects on powersupply performance far beyond those possible with even the mostaggressive designs in accordance with previously known technologies.Thus the advantages of the high efficiency of capacitors and the fasttransient capability of linear regulators (augmented by control throughimpedance matching) can be simultaneously achieved while obtainingoptimum use of capacitors which cannot be eliminated to advantage (tosupply the lower frequency transient power requirements) and overcomingmost of the undesired effects of parasitic inductance of capacitors andpower dissipation of linear regulators and provision of seamlessintegration of the VRM, packaging capacitors and linear regulator withfull compatibility with AVP, as will be further explained below.

For example, at the present time, a 2 MHZ switching frequency, f_(s), iscurrently considered to be aggressive switching regulator design. To usea switching regulator at the location proposed for connection of thelinear regulator in accordance with a preferred embodiment of theinvention would require a bandwidth, f_(c), of about 1.1 GHz or, usingthe 1/6 ratio discussed above, a switching frequency, f_(s), of about6,600 MHZ, greater than three thousand times the switching frequency ofswitching regulators of current aggressive design and which is notreasonably possible in view of the impedance due to ESL and therelationship of ESL and capacitance discussed above.

The key to implementing this principle of operation is to properlydesign the output impedance of the linear regulator 64 and the switchingregulator 60. One suitable technique of analysis and design sufficientto allow practice of the invention by those skilled in the art will bedescribed below in connection with FIGS. 8-12. Of course, many othersuitable techniques of circuit analysis and synthesis will becomeapparent to those skilled in the art in view of this example.

FIG. 8 shows a worst case transient waveform in which the maximum di/dtoccurs immediately after the conclusion of the maximum di/dt of theopposite sign. (This case is considered to be a “worst case” waveformsince any less aggressive waveform would be handled to a greater degreeby the packaging capacitors.) This waveform is scaled to the frequencyof the load circuit clock to which the switching frequency of theswitching regulator is also scaled. The waveform has two portions in ahalf cycle of the waveform: an initial portion with current increasingat the maximum slew rate of the power distribution system for about 10clock cycles or 20% of a half-cycle of the transient waveform and asecond portion with current rising at the maximum slew rate of theswitching regulator for about 40 clock cycles or about 80% of thehalf-cycle of the transient waveform. The second half-cycle is aninversion of the first portion. The upper trace of FIG. 9 shows the sameworst case waveform i_(o) of FIG. 8 with the current waveformi_(L)+i_(C) which can be supplied by the switching regulator and thepackaging and on-die capacitors and thus having a lower maximum slewrate superimposed thereon. As described above in connection with FIG. 7,the difference between these waveforms corresponds to the current andenergy supplied by the linear regulator 64 (FIG. 6) which is depicted byshaded portions of the lower trace of FIG. 9. The waveform of thecurrent supplied by the linear regulator 64 in this worst case exampleis thus seen to be a small triangular pulse of relatively low currentand duty cycle which limits the linear regulator power dissipation,alluded to above.

To obtain this response from the linear regulator, this time-domainwaveform may be placed in the frequency domain by suitabletransformation or simulation well-understood in the art as depicted inFIG. 10. After the desired current waveform in the frequency domain isachieved, the desired output impedance of the linear regulator and theswitching regulator can be easily derived using the equations:Z _(o) _(—) _(LR) /Z _(o) _(—) _(SR) =i _(SR) /i _(LR)  (1)and(Z _(o) _(—) _(LR) *Z _(o) _(—) _(LR))/(Z _(o) _(—) _(LR) +Z _(o) _(—)_(LR))=R _(droop)  (2)where the subscripts LR and SR denote the linear regulator and theswitching regulator, respectively, and R_(droop) is the droop resistanceor the amount the load/microprocessor voltage is required, by themanufacturer, to decrease as the load it represents increases in orderto maintain power dissipation and other operational parameters withindesign limits. Implementation of such a droop is referred to as adaptivevoltage positioning (AVP) alluded to above. These impedances of theswitching regulator and the linear regulator are graphically depicted asa function of frequency in FIGS. 11 and 12, respectively. The actualimpedances are too complex to be accurately modeled and practicallyrealized in a compensating circuit in a feedback path to control thelinear regulator as will be described below but may be approximated asillustrated by the dashed lines in FIGS. 11 and 12. (It should be notedin this regard that the apparent divergence of the dashed lines from theactual impedances above f_(c1) and f_(c2) which are two cornerfrequencies of the desired linear regulator output impedance, is notcritical or even relevant to the successful practice of the invention toachieve its meritorious effects but is convenient for simulation in viewof the likelihood of output impedance of the microprocessor will belower than the linear regulator at frequencies above 1 GHz.) It ispreferred, however, but not particularly critical to the practice of theinvention for the impedance of the linear regulator at frequenciesf_(c1) and f_(c2) to correspond to an impedance such that the impedanceof the linear regulator approximately matches the impedance of theswitching regulator at about 200 MHZ as indicated by point 110 of FIG.11. At a frequency point where the impedances match, the linearregulator and the switching regulator will be providing equivalentcurrents such that as the transient frequency increases the lowerimpedance of the linear regulator allows it to provide increased currentas the switching regulator and capacitors deliver less current as theESL of the capacitors increases. That is, as a practical matter, theswitching regulator bandwidth, by design, is f_(c1), at which point, thelinear regulator should preferably have a lower output impedance.Conversely, an impedance match at this frequency assures that the linearregulator cannot deliver significant current at lower transientfrequencies where the impedance of the switching regulator andcapacitors is less and adequate current can be supplied therefrom. Therespective impedances of the linear regulator, switching regulator andcapacitors as a function of frequency are illustrated in FIG. 13A (whichis substantially a superposition of FIGS. 11 and 12 with the capacitorimpedance overlaid thereon. It can also be seen from FIG. 13A that arelatively low output impedance is maintained by at least one (in orderof increasing frequency) of the switching regulator, the capacitors andthe linear regulator for a frequency range of over four decades beyondthe frequency at which the impedance of the switching regulator beginsto increase.

The design of the switching regulator with which the hybrid filter ofthe invention may be employed merely follows conventional practice andthe adaptive voltage positioning (AVP) design specified by the chipdesigner/manufacturer. The linear regulator principally comprises acompensator and a power amplifier. A preferred design for thecompensator 130 is illustrated in FIG. 13 which is simply an operationalamplifier having an input impedance network and a feedback impedancenetwork including RC connections as indicated, for example, by R1, R2,C1 and C2 having time constants appropriate to providing a low outputimpedance at the frequencies to be matched in the impedance curves ofFIGS. 11 and 12 and which assure that the power amplifier of the linearregulator of the hybrid filter will inject current into the power supplyconnection only when required power transients exceed a current slewrate of the switching voltage regulator and filter capacitors, howeverthe latter may be embodied. It should be noted in regard to thisfeedback circuit is connected to the sense point (corresponding to pointB in FIG. 3, upstream of the current injection point of the hybridfilter) and that no voltage reference is needed but only an input,{tilde over (v)}_(o), to the compensator which represents theperturbation of the voltage at the sensing point. Rather, theoperational amplifier with its feedback circuit should have sufficientbandwidth to ensure that it holds a −180° phase shift out to the cornerfrequency f_(c2) in FIG. 12. To achieve this, the operational amplifierwould need a gain-bandwidth product of at least 1/R_(droop)*f_(c2).Anything beyond this provides additional assurance that the propagationdelay is negligible.

Three exemplary preferred designs of the power amplifier circuit 140 areillustrated in FIG. 14. The preferred power amplifier structuresillustrated are of the well-understood push-pull configuration but thepower amplifier circuit is not at all critical to the practice of theinvention and many other suitable circuits will be evident to thoseskilled in the art. It should be noted, however, that the transistors ofthis amplifier are operated in the forward-active region, contrary tomost other attempts to provide enhanced transient response. Further, incombination with proper coupling to the compensator, this mode ofoperation can be performed without instability by proper design of theRC input circuits thereto and the output capacitor as described in theabove-incorporated article whereas instability was encountered in earlyattempts to utilize such a configuration.

FIG. 15 schematically illustrates an equivalent circuit of a preferredembodiment of the invention constructed as illustrated in FIG. 18 andcorresponding to the more generalized depiction of FIG. 6. The parasiticinductance of the linear regulator connection can be held to about 3 pHby locating the linear regulator adjacent to the processor on an organicland grid array (OLGA) above the microprocessor/IC socket as alluded toabove.

FIG. 16 is a schematic diagram of a preferred embodiment of the linearregulator in accordance with the invention formed by the combination ofcircuits 130, 140 of FIGS. 13 and 14 (including coupling circuits to PNPand NPN transistors for push-pull operation, depicted as amplifier 140′having an input, {tilde over (v)}_(o), derived from sense point B inFIG. 15 which was used for simulation of the operation of the inventionand FIG. 17 graphically illustrates the results thereof. (It should alsobe appreciated that the voltage at point B is also the output voltagefrom the first stage switching regulator as seen near the on-diecapacitor. As pointed out above, however, {tilde over (v)}_(o)represents the perturbation at the sensing point and does not includethe DC, or average, voltage but does represent disturbances to theswitching voltage.) The simulated current waveforms closely approximatethe theoretical waveforms discussed above in connection with FIGS. 7, 9and 10, particularly i_(LR), the current contribution of the linearregulator, confirming the low current and low duty cycle which limitpower dissipation of the linear regulator.

Also, in accordance with the results of this simulation, it is seen thatthe output/packaging capacitance can be reduced by over 60% and ESL canbe ten times larger in comparison with the prior art; indicating thatcapacitance requirement for the capacitor (e.g. the packaging capacitor)are not stringent in the context of the hybrid filter in accordance withthe invention. Further, because the linear regulator functions as acurrent source, it is not sensitive to the parasitic inductance of thepath between the linear regulator output and the microprocessor or otherload.

The concept of the hybrid filter using a linear regulator can beextended to a point-of-load converter since the circuits of FIGS. 13 and14 are relatively simple and of few components and thus may be placeddirectly on the chip or die with the circuits which they supply. Thisapplication of the invention may be of particular advantage intelecommunications devices to dramatically reduce the number ofcapacitors and total output capacitance required.

In view of the foregoing, it is seen that the invention provides asolution to more stringent power supply and distribution networkrequirements including adaptive voltage positioning as voltage isreduced, current increased and extreme clock rates are employed. Ratherthan increasing required capacitance to obtain adequate regulation whichcompromises slew rate of switching regulator power supplies, theinvention allows the capacitance to be drastically and dramaticallyreduced while accommodating unavoidable parasitic impedances whichbecome significant at high frequencies. The linear regulator functionsas a current source and can be placed close to the point-of-load tolimit parasitic inductance while power dissipation thereof is limited bylow current requirements and low duty-cycle operation such that theoverall power supply in accordance with the invention is notsignificantly lees than the efficiency of the switching regulator.Filtering requirements may be easily met by reduced capacitance valueswhich may be readily and conveniently provided. The performance of theinvention for meeting transient load changes is many times and severalorders of magnitude beyond the capability of the most aggressive designsin accordance with known switching regulator technology.

While the invention has been described in terms of a single preferredembodiment, those skilled in the art will recognize that the inventioncan be practiced with modification within the spirit and scope of theappended claims.

1. A method for reducing capacitance required for filtering output of avoltage regulator and/or increasing effective slew rate of said voltageregulator, said method comprising steps of setting bandwidth of saidvoltage regulator in accordance with a clock frequency of a load circuitreceiving power from said voltage regulator, injecting current into apower supply connection when required power transients exceed a currentslew rate of said voltage regulator by matching impedance of a currentinjecting circuit for injecting current with an output impedance of saidvoltage regulator.
 2. A method as recited in claim 1, including afurther step of determining when required power transients exceed acurrent slew rate of said voltage regulator and controlling said step ofinjecting current by said impedance of said current injecting circuitfor performing said injecting step.
 3. A method as recited in claim 1,wherein said bandwidth of said voltage regulator is set at approximately650 KHz.
 4. A method as recited in claim 1, including a further step ofsetting a desired adaptive voltage positioning (AVP) of said voltageregulator.
 5. A method as recited in claim 1, including a further stepof filtering output of said voltage regulator only with packagingcapacitors and/or on-die capacitors.
 6. A method as recited in claim 1,wherein said step of injecting current is performed at a point of load.